InGaP/GaAs HBT for 28 V operation is identical to common 5 V counterpart in its epitaxial layers except the collector thickness. Increasing collector thickness to 3 mm enables the base-collector breakdown voltage to go more than 60 V. Likewise, monolithic microwave integrated circuit (MMIC) for 28 V HBT is achieved with two interconnection metal layers, MIM capacitor, thin-film resistor, spiral inductor, and through substrate via holes. The substrate is thinned to 4 mils.

As a result, BV_cbo of 70 V is achieved with BV_ceo over 30 V. With 28 V as the bias voltage, the collector voltage in RF operation will swing much above 28 V and exceed the BV_ceo. Such condition does not present any concern to the device operation or its reliability^[8].

Power HBT, like other semiconductor technology, is made of multiple small devices strung in parallel. Thermal resistance design is the first task for any power device. Sufficient spreading of the active HBT fingers across the IC die is done in the conventional MMIC approach. Bipolar transistor is known to require ballasting since V_be has a negative temperature coefficient.

The individual HBT finger size is balanced between the RF performance and thermal resistance. Multiple HBT fingers are linked into the basic building block. The building block in the present design delivers around 2 W RF power at 2 GHz. Each building block has a MMIC prematch circuit.

The bias circuit for the 28 V power HBT is implemented on the same chip through the current mirror approach. Excellent thermal stability is achieved: less than 9% change in the quiescent current is achieved over -40 °C to +85 °C.

Linearity improvement: The driver stage for the power amplifier chain is often biased toward class A in order to provide the needed linearity. This approach sacrifices the operation efficiency. It was found that a low-frequency low-source impedance matching improves the linearity in near-class B operation^[4,6]. At the dc side of the choke, a 6 µF shunt capacitor is used to provide a low impedance at the modulation frequency and 5 dB improvement in the IM3 is observed. The improvement comes from the elimination of the low-frequency component (ω₁-ω²) at the input, which if existing will mix with ω₁ and ω₂ to generate the third-order distortion^[6].

Further improvement on the linearity in class AB or class B operations were achieved via a dynamic bias circuit. The major non-linearity in the bipolar transistor is found to come from the exponential I-V relationship. Following the previously reported analysis^[11], HBTs are found to have similar behavior as LDMOS in the third-order derivative of the function I_c(V_in) as shown in Figure 2.

The curve of I_c-V_in along the load line follows the exponential relationship with the ballasting resistor effect. The quiescent bias point is at point I. For conventional class B operation, the dc average voltage will remain at point I regardless of the RF swing. With the dynamic bias circuit, the time average bias point is lifted up to conditions II then III as the input power level is increased. The RF voltage avoids swinging into the peaking portion of the g_m (V_in) curve as the bias point is lifted by the dynamic bias circuit. Thus, improving the linearity in near-class B operation. Figure 2 shows the analysis of the third-order intermodulation distortion (IMD3) contributed by the transconductance non-linearity in the bipolar transistor.

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